Posted on

PB2021.10 Study of CDM Relay Pogo-Contact First CDM and CC-TLP Pulses

Download PDF – Study of CDM Relay Pogo-Contact First CDM and CC-TLP Pulses

Abstract – The recently proposed RP-CCDM testing method is tested alongside the more established CDM and CC-TLP methods on two test ICs to assess the RP-CCDM’s efficacy as an alternate test method. High bandwidth analysis of field induced CDM spark events is presented to evaluate higher frequency components of the current.

I. Introduction

A CDM event occurs when the pin of a charged device approaches an external metal object such that the potential difference exceeds the breakdown voltage of the air gap between them. CDM is one of the most common ESD threats in modern manufacturing and usage environments. The current industry standard testing method is the Field Induced CDM (CDM) method governed by the ANSI/ESDA/JEDEC JS-002 standard [1]. Current CDM testers are plagued with repeatability issues due to the variable spark resistance of the air discharge making the practice difficult to standardize [1, 2]. Additionally, decreasing CDM testing voltages increase the pulse-to-pulse variability of the air discharge, causing concern over the ability to meaningfully classify devices at lower voltages [3]. This is becoming increasingly problematic as the necessity for classification at lower levels is becoming greater with advances in IC technology [2]. Several contact-first CDM methods where a CDM current is induced into the pin through a more controllable method, such as the CC-TLP, LICCDM, and RP-CCDM, have been introduced in an attempt to solve the issue of pulse variability [4, 5, 6].

This work presents a correlation study between the CDM, RP-CCDM and CC-TLP testers on two devices with well-known CDM failure levels and failure mechanisms. The devices are subjected to a series of RP-CCDM and CC-TLP tests with multiple risetimes. The study aims to evaluate the efficacy of the RP-CCDM method and gain insight into the correlation of RP-CCDM and CC-TLP to better understand their potential as replacements for CDM susceptibility testing.

II. CDM and CC-TLP Tester Configurations

A. RP-CCDM Tester

The RP-CCDM, or Relay Pogo-Contact First CDM, is a design of the CDM discharge head that allows the use of a repeatable relay discharge while largely preserving the design parameters of the JS-002 standard. Figure 1 shows a cross section of the RP-CCDM head.

As shown in Figure 1, the RP-CCDM uses the field charging method and a similar discharge path to the one specified in the JS-002 standard. To charge the device, the pogo pin of the RP-CCDM ground plane is lowered to contact the DUT pin, then the field plate is brought to the specified charge voltage. To discharge, the reed switch is closed and the current primarily flows up the pogo pin, through a high bandwidth 1 Ω resistor, and returns via the ground to field plate and DUT capacitances. A more detailed testing procedure for the RP-CCDM is presented in [6].

The RP-CCDM prototype head used for stressing devices in this study was found to adhere to the discharge waveform parameters specified in the JS-002 standard. The RP-CCDM prototype was also found to have a strong correlation in peak currents to the peak values measured with a CDM head across a range of coins with varying CDUT values.

B. ACRP-CCDM Test Setup

The RP-CCDM and CDM test are different in two key aspects that could be the source of correlation issues. First, the RP-CCDM spark takes place in a controlled environment free of air. This difference results in a much more reproducible spark environment that reduces the variability of the peak current when compared to air discharge CDM and enables testing below 100 V [6]. The altered spark environment may also influence the risetime and result in less damping of the system due to a smaller series spark resistance value. Second, the geometry of the RP-CCDM head is different from that of the CDM to accommodate the reed switch. Specifically, the pogo structure is no longer uniform in thickness and longer than in CDM, which will affect the inductance of the discharge path. To further investigate the role of these two variables the “Always Closed” RP-CCDM (ACRP-CCDM) test method was implemented. This test method uses the existing RP-CCDM test setup but alters the testing procedure to close the reed switch in the pogo pin prior to descent such that a CDM-like air spark is created. This method removes the reed switch spark variable and reveals differences between CDM and RP-CCDM based on the test head geometry alone.

To evaluate differences in the rising edge, or any other part of the current waveform, RP-CCDM, ACRP-CCDM, and CDM tests were performed using a 23 GHz bandwidth Tektronix MSO72304DX oscilloscope and a low loss measurement chain. The insertion loss of the measurement chain, shown in Figure 2, was less than 2 dB up to the 23 GHz measurement bandwidth of the oscilloscope. The same model of disk resistor was used for the construction of both heads with the resistive sheet facing away from the pogo-pin, this type of disk resistor has been characterized up to 26.5 GHz in [7]. The DC resistance value was used for current measurement scaling.

Figure 3 displays the five highest peak current pulses captured from a set of fifty discharges on a small JS-002 verification module at TC500. These pulses represent the lowest spark impedance discharges for each tester. The initial rising edge of the CDM and RP-CCDM pulses is observed to be similar, although the CDM rising edge reaches a higher peak in the first 50 ps and the RP-CCDM reaches a higher peak in the first 80 ps. The ACRPCCDM current measurement has a rising edge more similar to that of the CDM pulse, possibly indicating a relationship between the rising edge and the air spark event, whereas for the rest of the measurement duration the ACRP-CCDM pulse matches the curve of the RP-CCDM current indicating a relationship with the test head geometry. The risetime of the RP-CCDM does not exhibit any characteristics that indicate a faster risetime than its air spark counterparts. The RP-CCDM slew rate may be faster on average, however, due to degradation in the CDM slew rate during discharges with high series spark resistance. Figure 5 displays the frequency spectrum of each pulse shown in Figure 3.

In [7], it was observed that the high frequency content seen in the CDM and RP-CCDM spectrum is likely related to the pogo pin length and the disk resistor. As shown in Figure 5, the high frequency content observed in the RP-CCDM spectrum is quite similar to the content seen in the ACRP-CCDM spectrum, indicating that the RP-CCDM relay spark does not introduce any unwanted resonances. Whether the current measurements in Figure 3 and the structure resonances seen in Figure 5 are indicating the true current magnitude that is traveling though the IC pin during a CDM stress is not yet fully understood but is currently being investigated. The presence of the high frequency current components that occur in field induced CDM events could be an important factor in determining the efficacy of alternate test methods on sensitive ICs.

C. CC-TLP Test Setup

The CC-TLP method utilizes a VF-TLP pulse into to a single IC pin and a capacitively coupled return path to create a CDM equivalent stress [4]. CC-TLP is a contact-mode tester and as such has the advantage of having much lower pulse to pulse variability than a standard CDM test with an air spark. The CC-TLP method also allows correlation between peak current levels and pulse width settings to CDM stress parameters which can be used to evaluate CDM type failure modes [8, 9]. In this study, two CC-TLP setups were used for testing. The first tester used to perform testing at IFX, uses a circular ground disk with a diameter of 5 cm. The length of the probe tip is adjusted to 0.3 mm [10]. A TLP system with 1 ns pulse width and 100 ps risetime is connected as excitation source. The second CC-TLP tester, manufactured by and used to perform testing at ESDEMC, uses a test head with a square, rather than circular, ground plate the same size as a JS-002 CDM tester’s ground plate. The length of the probe tip is adjusted to 0.5 mm. The bandwidth of both CC-TLP measurement chains were limited by the pick-tee cutoff at approximately 10 GHz. Both testers integrate S-parameter compensation of the measurement chain in software. The same pulse parameters were used for both testers. A “dummy” device was used to determine the charging voltage necessary to achieve the desired peak current on each pin of each device tested.

III. Correlation Study

A. Device Information

Two different test devices are used for the correlation study of RP-CCDM, ACRP-CCDM, CC-TLP and CDM in terms of the failure level. Both devices are designed in 130 nm CMOS technology. Device A is packaged in a TSSOP, device B as eWLB. The expected failure mode is the excess of a critical potential level leading to gate oxide damage at one or more transistors. The failure mechanism of device A is a GOX failure triggered by a cross-domain issue and is located at an internal interface in the core. Device B shows several GOX failures and increased IDDQ currents after the damage occurring at the transistors on the edge of the digital core.

B. Transmission Line Connected Pins

As observed in measurements on the JS-002 verification modules, the peak current of the RP-CCDM and CDM testers was measured to be nearly identical on a reference connected pin of the device as seen in Figure 6. However, this relation was found to not hold across all pins on device B.

The most sensitive pins on device B are high-speed I/O pins connected to the die via impedance controlled, transmission line (TL), traces, such as shown in Figure 7.

In Figure 8 two curves of a TDR measurement in a CC-TLP setup are shown. The probe tip of the CC-TLP test head is connected to the chip, so that the current path to the die can be analyzed such as shown in [11]. Once the current wave reaches the die, the capacitance formed by the package and die to the ground disk of the test head is charged and marks the end of the transmission path. This method enables to measure delay times of single pins. In case of a pin with a corresponding ball in the direct vicinity of the die pad the delay due to the transmission path can be neglected (Fig. 8, green curve). For the second pin the connected transmission line forms a stable impedance for about 120 ps (Fig. 8, blue curve).

The stress observed on these pins is paramount to determining the stress that the device can withstand. However, due to the high variability of air discharge CDM testing, it can be difficult to evaluate the difference between testers on these pins after a typical CDM validation test with only three discharges per polarity.

C. Peak Current Comparison of Pin Types

To evaluate these pins more accurately, a 100-pulse test was performed by stressing a reference connected pin and a TL pin 100 times each. All testers were tested at a 500 V field charge voltage to prevent any spark differences that may occur if the JS-002 voltage factor method was incorporated. The pogo pins and dielectric surface were cleaned with isopropyl alcohol prior to testing and the testing was performed below 10% relative humidity to minimize peak variation due to the air spark. Due to the charge distribution over the whole chip and high current levels with ESD devices operating in the on-state, no difference is expected in the discharge waveforms from before failure to after failure of the device. An additional CDM tester head with a replaceable pogo pin was incorporated to evaluate the influence of various pogo lengths on the observed discharge peak. This new test head was used alongside the RP-CCDM and CDM heads used in all device testing. The CDM head used for device testing is indicated as “large diameter” since it has a diameter of 1.5 mm compared to the new test head’s 0.4 mm pogo diameter. Table 2 displays the maximum peak current measured within the 100-pulse test for each setup and the relative percentage of the TL pin peak current to the reference connected pin peak current.

As can be seen in Table 2, there is a correlation between the length of the pogo pin and the discharge current ratio between the TL connected pin and the GND pin. For a given pre-charge voltage (with voltage factors applied such that the RP-CCDM and CDM peak currents match on JS-002 verification modules) where a CDM stress induces 5 A peak current on the studied TL pin, the RP-CCDM stress will induce a 5.7 A peak current. This additional current can be clearly seen in the measured currents shown in Figure 9, which was taken during a qualification test of device B. The cause of the increased peak current of RPCCDM with respect to CDM for the same pre-charge voltage on TL pins can be explained by the input impedance of the CDM head. Figure 10 displays the input impedance of the CDM heads used in Table 2 as measured in [6], denoted as ZDUT. Due to higher input impedance in the 600 MHz to 2 GHz range, where the majority of the spectral content is contained (Fig. 5), a larger reflection back into the device trace will occur in the case of the RP-CCDM. As discussed in [12, 13], the stress the die experiences is a result of the reflected pulse off this discontinuity. As a result, the stress a TL connected I/O receives will be greater for a CDM head with a higher input impedance, given a matching peak amplitude on a calibration coin.

D. Simulation of Transmission Line Pins using ZDUT

To determine if the measured ZDUT values alone could predict the difference, a simulation using the measured ZDUT data was done in ADS by Keysight. A simplified circuit model, developed in [12], was adopted to represent a CDM event on a TL pin where the current must travel through the impedance-controlled trace and encounter an impedance discontinuity at the pin/pogo-pin interface.

As can be seen in Figure 13, the ADS simulation produces the same relationship in measured current as the measured data. The current measured using the measured RP-CCDM ZDUT to represent the test head structure results in a higher current measurement on the TL connected pin. Figure 14 shows the current measured on the die side of the transmission line and indicates an increased current peak with the RP-CCDM head. Further development with a less simplified model of the TL pin case is necessary to determine the precise stress difference that occurs due to this effect.

IV. Failure Analysis Results

A. Device A Failure Results

Device A was tested using RP-CCDM, CDM, and CC-TLP testers. To test if that the failure levels of device A are risetime dependent, CC-TLP tests were performed using 100 ps and 300 ps risetimes. The determined failure levels of the device A are shown in Table 3 and Table 4. The devices were stressed at increasing test conditions until failure. The recorded withstand current indicates the highest current that all tested devices were able to withstand. Failure analysis on device A was carried out via DC leakage testing.

Table 3 and Table 4 display the failure levels for the device A. The RP-CCDM and CDM were observed to correlate well on this device, inducing failures at TC750. Additionally, CC-TLP setups at ESDEMC and IFX both induced failure at 8 A. Figure 15 displays a comparison of each tester’s discharge waveforms recorded at the failure threshold of device A. As seen in Figure 16, a clear jump in the leakage current is evident for each device stressed at its failure threshold.

B. Device B Failure Results

Device B was tested with the RP-CCDM and ACRP-CCDM at test conditions of 400 V, 500 V, and 625 V. Additionally, tests with CC-TLP setups were performed at 4 A, 5 A, and 6 A with a 100 ps risetime and 6 A, 7 A, and 8 A with a 300 ps risetime.

Failure analysis on device B is carried out via DC leakage testing and verified via IDDQ testing. The failure is expected for pre-charging voltages higher than TC500 on CDM and located on the edge of the digital core. The IDDQ analysis measures the current consumption in the quiescent state on 200 vectors in the digital core.

The quiescent current into VDD for all vectors is below 100 uA in case of an unstressed reference device. Slight deviations in this range are traced back to normal variations. A significant increase of the current compared to the reference device indicates a gate-oxide failure on the edge of the digital core. Significant deviations between vectors in the IDDQ sweep are also an indication of failure on the edge of the digital core.

Table 5 and Table 6 display the failure levels for device B. Figure 17 – Figure 20 display the IDDQ readouts indicating the failure thresholds for each tester.

As shown in Table 5, the RP-CCDM induced failures below the TC500 withstand voltage of the device on a CDM tester. Figure 21 displays the measurement of the largest amplitude stress discharge for the CC-TLP, ACRP-CCDM, and RP-CCDM stress sets at 5 A, TC500, and TC500, respectively.

As shown in Figure 21, the ACRP-CCDM and RP-CCDM have similar pulse shapes and amplitudes on the sensitive I/O pin studied previously and produce failures at the same test condition. The RP-CCDM IDDQ sweep shows a more larger failure, which may be due to the more controlled spark environment and therefore slightly higher peak currents. Both the RP-CCDM and ACRP-CCDM testers produce increased amplitude pulses on TL pins when compared to the CDM tester due to the higher input impedance as discussed in detail in section III. The failure of both the ACRP-CCDM and RP-CCDM to correlate to the CDM at TC500 indicates that the geometry of the test head is likely responsible for the early failures on device B, rather than the spark event in the relay environment. The geometry contributes to a higher stress in the case of ACRP-CCDM and RP-CCDM due to the TL pin effect studied in section III.

As shown in Table 6, the peak current failure thresholds of both CC-TLP testers correlated well with each other by producing repeatable failures at 5 A peak current with a 100 ps risetime pulse and at 8 A with a 300 ps risetime pulse. It was observed during the correlation study that a failure at 5 A could not be reliably achieved until the risetime of the incident TLP pulse at the CC-TLP probe port was slightly decreased and approached 90 ps on the ESDEMC tester. This observation reasserts that the risetime of the stress is a critical stress parameter of device B and that the exact risetime of the incident TLP pulse is an important factor to consider when correlating CC-TLP systems.

V. Conclusions

In this study, the RP-CCDM test method was evaluated in comparison to the CDM and CC-TLP test methods using high bandwidth measurement and spectrum analysis, input impedance measurement and circuit modeling on TL pins, and device failure analysis on two devices.

It was shown that a longer pogo pin can induce a larger amplitude current discharge on a TL pin, given a matched peak current on a reference pin, due to the larger input impedance seen at the pogo-pin/device pin interface. This effect was replicated using a simplified model in ADS.

The RP-CCDM showed perfect correlation to CDM stress tests on one device, but consistently showed failures at a lower threshold on a second device. Although construction and excitation sources of both CC-TLP testers are different, they correlated well with the failures induced by the CDM and were helpful in indicating whether risetime was a critical stress parameter of the test devices.

The RP-CCDM was also used to perform air discharge tests, when the relay in the pogo-pin was closed during the entirely of the test and resulted in failure levels that matched RP-CCDM results. These results indicate that the RP-CCDM structure is likely responsible for the early failures on the second device due to the increased currents induced on TL pins. Analysis of the spectra between CDM (air spark) and RP-CCDM (relay spark) suggest that there is no significant increase in spectral content other than that caused by as more stable, low impedance spark. This observation suggests that the RP-CCDM may represent a “worst-case” air spark, that is the spark that would occur between two very clean surfaces at low relative humidity. More testing and statistical analysis of the RP-CCDM is necessary to make a conclusion about the “worst-case” analogy.

VI. Acknowledgments

The authors would like to thank Dr. Kai Esmark of Infineon Technologies AG for providing the testing devices, for his support regarding the failure analysis and for sharing his wide knowledge of ESD, and in particular, of CDM. The authors would also like to thank Michael Reardon of ESDEMC for his assistance and input during the CC-TLP testing portion of this study.

Posted on

PB2021.10 Analyzing the Influence of Imbalanced Two- or Three-Wire VHF LISN on Radiated Emissions from AC Cables

Download PDF – Analyzing the Influence of Imbalanced Two- or Three-Wire VHF LISN on Radiated Emissions from AC Cables 

Abstract—This article investigates using imbalanced two- and three-wire terminations for ac main cables, as suggested by the standard group. These terminations provide the basis for a new line impedance stabilization network (LISN) whose objective is to improve test repeatability between labs while also providing better estimation of real-world emissions. Standard balanced LISNs do not reproduce the imbalanced terminations seen in practice. An imbalanced two- or three-wire very high-frequency LISN was built, which can handle up to 15 A on each line. The LISN operates from 30 to 200 MHz and provides greater than 50-dB isolation between the input and output. The imbalanced termination allows the device to create a specified degree of conversion from differential-mode to common-mode current, which can increase radiated emissions. This conversion was evaluated to be as high as 12 dB in measurements of a power line communication device. 3-D full-wave simulations of two- and three-wire applications demonstrate that the radiated emissions from the prototype LISN and the ideal imbalanced termination are nearly equal. The new LISN was further evaluated to show promise for improving measurement reproducibility, reducing the standard compliance uncertainty by 6 dB in this study, from 15.5 dB in CISPR 16-4-1 to 9.5 dB with the LISN.

Index Terms—Common-mode impedance, radiated emissions, termination device, very high-frequency line impedance stabilization network (VHF LISN).


Common mode conducted emissions are typically measured using a line impedance stabilization network (LISN) [1]–[9]. An LISN is a filtering device providing a) isolation of the device under test (DUT) from ac power lines and related radio frequency (RF) disturbances, b) a well-defined reference impedance at the LISN DUT port; and c) the necessary power to the DUT. Standard LISNs use a balanced termination structure [3]–[9]. Round-robin testing for radiated emissions show that the average emissions differ by 4 dB when using a traditional balanced LISN (see Fig. 1) [3], [4], but those deviations in measurements were as large as +18/−10 dB when the DUT was plugged directly into the building mains [3], [4]. Although a goal of the balanced very high-frequency (VHF) LISN is to reduce variations among tests [1]–[9], a balanced termination is rarely available in practice, so real-world emissions may be higher than seen in many standardized measurements.

The LISN forms an impedance between the wires of the power cable and the chamber ground and, thus, can be used as a common-mode absorption device above 30 MHz. If the common-mode impedance of the LISN is between 50 and 300 Ω, most of the common-mode resonances of the power cable will be suppressed. Suppressing these resonances reduces the dependence of the emissions on the power cable routing, the specific impedance of the chamber’s power connection, and the length of the power cord. While reducing this dependence is attractive for minimizing chamber to chamber variations, it also hides an important effect that causes radiation. In real installations, the differential-mode noise current is often larger than the common-mode noise current in a power cable, and real installations will have asymmetric common-mode impedances [10]–[12]. This asymmetry will convert differential-mode current into common-mode current, which can radiate. To mimic this effect, a defined asymmetry can be introduced into the VHF LISN.

Using an imbalanced termination will increase the common-mode emissions by about 10 to 15 dB up to 200 MHz (depending on the DUT) compared to using a balanced termination [11]. About 10-dB higher emissions were also reported for an imbalanced two-wire LISN compare to a balanced LISN over 0.5 to 30 MHz in [11]. To address the differential- to common-mode current conversion in a VHF LISN (30 to 200 MHz), the termination impedance should have a controlled imbalance to provide a defined degree of conversion from differential to common-mode current. The standard group introduced an imbalanced termination for this purpose and suggested that an LISN with similar performance should be created for the 30- to 200-MHz frequency range [10].

In this article, an imbalanced LISN with the characteristics suggested by the works in [10]–[12] was designed, built, and analyzed to serve as the termination of the mains power during radiated EMI CISPR16/CISPR 35 testing [1]. This LISN was designed in Section II to supply power for imbalanced two- or three-wire measurements up to 15 A over a frequency range from 30 to 200 MHz. In Section III, a 3-D full-wave simulation with both differential- and common-mode excitations representing the DUT was used to illustrate the effect of the VHF LISN on radiated emissions on a typical test setup for two- or three-wire application. A study of differential- to common-mode current conversion was performed to verify the conversion ratio for an ideal imbalanced termination. In a real test setup, the differential to common-mode current conversion is geometry dependent but not a fixed number. Measurements of a pair of power line communication devices were performed in Section IV to validate the performance of the LISN in a typical test setup and to demonstrate the level of differential- to common-mode current conversion for a real-word application. Discussions are presented in Section V. Finally, Section VI concludes this article.


The following section will introduce the imbalanced two-wire and three-wire terminations suggested by the working group and the literature [10]–[12], provide evidence of the suitability of the proposed impedances, show the characteristics of the actual LISN that was built to use these terminations, and give the value of differential- to common-mode conversion for a practical test setup connected to the imbalanced two wire termination.

A. Imbalanced Two- or Three-Wire Termination

The imbalanced two- and three-wire terminations proposed by the works in [10]–[12] are shown in Fig. 2(a) and (b), respectively. The termination characteristics for two-wire applications were specified as follows.

1) Common-mode impedance ZCM of 150 Ω ± 10% from 30 to 200 MHz, which defines the impedance between the line (L) and neutral (N) wires and the ground-plane, when the line and neutral wires are shorted together.

2) Differential-mode impedance ZDM of 100 Ω ± 10% from 30 to 200 MHz, which defines the impedance between the line and neutral wires when the neutral wire is shorted to the ground plane.

3) Impedance between the line and ground plane of 250 Ω ± 20% from 30 to 200 MHz.

The termination characteristics for three-wire applications were specified as follows.

1) Common-mode impedance ZCM of 90 Ω ± 10% from 30 to 200 MHz, which defines the impedance between the line, neutral, and protective earth (PE) wires (shorted together) and the ground plane.

2) Differential-mode impedance ZDM of 100 Ω ± 10% from 30 to 200 MHz, which defines the impedance between the line wire and the neutral and PE wires, when the neutral and PE are shorted to the ground plane.

3) Tertiary-mode impedance ZTM of 60 Ω ± 10% from 30 to 200 MHz, which defines the impedance between the line and neutral wires shorted together, and the PE wire and ground plane shorted together.

The working group [10]–[12] chose a 150-Ω two-wire common-mode impedance to suppress standing waves in the DUT power. The geometry of a power cable does not allow for easily assigning a common-mode impedance, as the wave structure deviates strongly from a TEM mode wave, but it has been shown that termination impedances in the range of 50 to 200 Ω suppress standing waves well [13]. Although not exact, using a common-mode impedance of about 150 Ω [see Fig. 2(a)] will provide results similar to those observed in [13] and minimal common-mode resonances will occur [13]–[15].

A 90-Ω termination was selected for the slightly thicker three-wire cables [10]. The 100-Ω differential impedance was chosen to mimic the impedance of a typical transmission line (TL) formed by the line and neutral wires [10]. To provide evidence for the validity of this selection, 48 different power cables were measured using a time-domain reflectometer (TDR) (see Fig. 3). Eleven were two wire cables, and 37 were three wire cables.

The impedance value was recorded between the short discontinuity at the beginning of the coax to cable transition and the end of the cable, which was left open. As shown in Fig. 3, a small adapter was made to connect the coax cable to the power cord, which causes a short discontinuity but does not affect the TDR measurement, as we record the differential impedance between the two lines after that discontinuity. The length of the cables is not important as we look at the differential-mode impedance provided by two wires (L and N).

The distribution of the characteristic impedances is shown in Fig. 4. The distribution shows the typical value of the differential impedance is close to 100 Ω, as suggested by the working group [10]–[12]. In the tertiary mode, the line and neutral wires are shorted and driven against the PE and ground (shorted). The line and neutral wires together as a larger “signal” conductor compared to the differential mode case, so the TL impedance is expected to be smaller. A value of ZTM = 60 Ω was chosen for this reason [10].

B. Prototype Imbalanced Two- or Three-Wire LISN

An imbalanced LISN was designed to meet the specifications of the working group. Additional requirements for the designed LISN are listed in Table I. Fig. 5 shows the LISN’s circuit diagram. The capacitor inductor networks in Fig. 5 (L1, C3, L4, and L2, C4, L5) were built to provide the required 50 dB of isolation of the DUT from the mains network at higher frequencies and provide a good connection to the building mains at 50 to 60 Hz. The capacitors C1 and C2 were made sufficiently small to isolate the line and neutral wires from each other and the PE at 50 to 60 Hz, but to allow resistors (R1, R2, and R3) to define the termination from 30 to 200 MHz. While the schematic in Fig. 5 should meet the specifications of the working group, validation is required since parasitics could alter the actual impedances looking into the LISN.

The six impedances specified by the working group could not be measured directly, since they require a floating measurement and different parasitics will be involved in each measurement. To determine the impedances, a combination of measurement and postsimulation was used. Three-port S-parameter measurements were made looking into each terminal of the LISN. A test fixture (see Fig. 6) was built to connect the LISN to the vector network analyzer for the three port S-parameter measurement. The measured S-parameter matrix was imported into advance design system (ADS) [16] to calculate the six specified impedances. The measured impedances and those suggested by the working group are shown in Figs. 7, 8, 9, and 10. The magnitudes of the measured impedances are all within the ranges proposed by the working group [10]–[12]. While the working group does not explicitly specify the phase, the measured phase is within about 30◦ of the ideal 0◦ phase for a resistive termination. More important than the variations in the actual impedance is its impact on the radiated fields. It will be shown in Section III that up to 30◦ variation in phase plus 10% variation in the magnitude of the LISN impedance will cause less than ± 3 dB variation in the radiated emissions, which is acceptable in EMC applications. These results suggest a ± 10% change in magnitude and ± 30◦ in phase is an appropriate limit for defining the impedance of an imbalanced VHF LISN.

C. Impact of Imbalance on Differential- to Common-Mode Conversion

An LISN is a filtering device providing the following:

1) isolation of the DUT from ac power lines and related RF disturbances;

2) a well-defined reference impedance at the LISN DUT port, which is isolated from the main network;

3) the necessary power supply to the DUT.

Looking from the DUT side, the common-mode impedance in the two-wire imbalanced LISN investigated is 150 Ω [see Fig. 2(a) and 13]. However, the common-mode impedance of a balanced LISN is about 50 Ω as seen from the DUT side. For both LISNs, the differential-mode impedance in the two-wire setup is 100 Ω [11], [12].

Testing without an imbalanced termination misses the important differential- to common-mode current conversion that may occur when the product is connected to a real ac mains network. The impact of the imbalanced LISN on differential to common-mode conversion was demonstrated using a 3-D full-wave model of a typical test setup. The model shown in Fig. 11 illustrates a typical radiation test setup [1]. The DUT is represented with a solid metal box (30 cm × 10 cm × 30 cm) located 1 m above an infinite ground plane. The box is connected to a 1.5 m power cable. The wires in the cable were driven with a 1-V differential source with zero-output impedance, as shown in Figs. 12 and 13. The sources were connected to the DUT chassis with a low impedance (10 Ω). This impedance is intended to represent a poor connection of a shield to the chassis. This connection was compared to a 1-Ω connection, which showed that this selection of the connection impedance does not significantly influence the conclusions drawn from the simulation.

A typical power cord geometry was used having a 1.62-mm wire diameter, a 0.89-mm-thick PVC insulation, and a PVC jacket with a diameter of 9.5 mm. The metal to metal distance between the wires was 2.35 mm. This distance forms a TL of about 80 Ω between the line (L) and neutral (N).

Circuit schematics of the full-wave structure with balanced and imbalanced terminations are shown in Figs. 12 and 13, respectively. The balanced or imbalanced terminations were chosen similar to those recommended for the LISN (see Fig. 2). The capacitance between the DUT enclosure and ground is about C ≈ 50 pF, depending on the size of the DUT. This capacitance creates an impedance of about 15 to 100 Ω from 30 to 200 MHz, which is on the order of the 150-Ω common-mode termination RCM, so that nonnegligible common-mode current could flow. Simple models without TLs and an estimated value of the coupling capacitance between the DUT and ground can be evaluated in [11]. Each wire is driven with an identical 0.5-V voltage source if opposite sign is creating a 1-V differential-mode source, which drives differential-mode current. As indicated in Fig. 12, there is no common-mode current when using a balanced termination, whereas the imbalanced termination will cause differential- to common-mode conversion of current (see Fig. 13), which will increase radiated emission. According to the works in [10]–[12], real installations have imbalances, which should be reproduced by the EMC test setup.

To calculate the amount of differential- to common-mode conversion, the full-wave model in Fig. 11 was simulated with an imbalanced two-wire termination (see Fig. 13) over a frequency range from 30 to 200 MHz, as shown in Fig. 14. The differential-mode current reduces with frequency since the 80-Ω TL is terminated into 100 Ω. Of course, for a different cable geometry, the differential-current (blue curve in Fig. 14) might be slightly different but that should not affect the current (< 10% variation) as long as the differential-mode impedance is between 80 to 100 Ω.

The common-mode current is affected by the structural length and fluctuates around 1 mA. The differential- to common-mode conversion can be measured by the ratio of the incident differential-mode power to the resulting power in the common mode as

The differential- to common-mode conversion ratio has been reported to be about −10 to −15 dB for an imbalanced CDNE-M [11]. The actual conversion ratio obtained through full-wave simulation, however, is somewhat geometry dependent and varies from −9 to −25 dB, as shown in Fig. 14. For a threewire application [see Fig. 2(b)], the maximum differential- to common-mode conversion was similarly evaluated to be about −19 dB. The conversion is smaller for three wires than two, because current will return not only via the ground plane, but also in the PE wire. The portion that returns in the PE wire does not contribute to the common-mode current.


To demonstrate the impact of the imbalanced LISN on radiated emissions, emissions were simulated using the fullwave model shown in Fig. 11. As shown in Section II-B, the impedances of the actual LISN vary about the ideal values due to unintended parasitics. Simulations were performed using the ideal LISN termination impedances and the realized values, as well as short and open terminations to show performance at the extremes. Table II summarizes all the settings used in the computer simulation technology (CST) Studio Suite. Simulation with the realized termination impedances was accomplished using the cosimulation feature of CST with the measured Sparameters. This feature combines the 3-D full-wave simulation with a circuit simulation or measurement [18] to simultaneously solve for the EM fields and the circuit characteristics. To ensure correct cosimulation, the following simulation procedure was used.

1) The setup shown in Fig. 11 was simulated with the ideal termination impedances represented as lumped elements [see Fig. 15(a)]. This simulation demonstrates the ideal performance of the imbalanced LISN and provides a reference for validation of the cosimulation approach.

2) A second simulation was performed using the ideal terminations with cosimulation [see Fig. 15(b)], where the excitation and the loads were replaced with S-parameter ports. Using the CST design studio cosimulation feature shown in Fig. 16, the excitation was connected to the source port and an S-parameter block representing the loads was connected to the load ports. An S-parameter block evaluated for ideal terminations [see Fig. 17(a)] was used to verify the procedure against the reference result from step 1.

3) After successful validation, the S-parameter matrix measured on the prototype was used to calculate the radiated emissions.

Radiated emissions were also evaluated for short and open terminations, when all the impedances on the termination side were replaced with open or short. The radiated emission was evaluated for all cases at a distance of 10 m. Results were evaluated for both common- and differential-mode excitations.

A. Common-Mode Excitation With Imbalanced VHF LISN

The schematic for the simulation setups with a common-mode excitation and ideal imbalanced two- and three-wire terminations are shown in Fig. 18. The maximum simulated far-field radiation for the three-wire setup with the studied terminations is shown in Fig. 19.

shown in Fig. 19. Strong resonances were observed for open and short terminations. The first peak is around 50 MHz, which is below the quarter wavelength frequency for a 1.5-m wire length, because the 50-pF capacitance between the DUT enclosure and the surroundings. At resonances, the radiation can exceed the radiation from the ideal termination by up to 15 dB. The 90-Ω commonmode termination impedance effectively damps resonances and in that regard seems reasonable with respect to repeatability. These results are in agreement with the previous study on the effect of common-mode impedance on radiation [2]. As shown in Fig. 19, the termination impedance is important at lower frequencies, but is not as significant at higher frequencies due to the increasing electrical length of the wire and the impact of damping due to radiation.

The black and blue curves in Fig. 19 show the results using the cosimulation technique and using standard simulations with ideal terminations. The two curves are nearly identical (< 1 dB difference). The green curve shows the radiated emissions using the measured termination S-parameters from the prototype. Comparing the blue and green curves, the radiated emissions from the real LISN is nearly equal to the radiation seen for the ideal termination case. Similar results were seen for the two-wire common-mode case [e.g., using terminations as in Fig. 18(a) and (b)]. The radiated emissions from the prototype LISN are close to the emissions from an “ideal” imbalanced LISN with less than 2-dB error up to 200 MHz (red curve in Fig. 19).

B. Differential-Mode Excitation With Imbalanced VHF LISN

Fig. 20 shows the differential-mode source excitation and the imbalanced termination in two- and three-wire setups. For both setups, the DM excitation was 1 V with 0-Ω output impedance. A low impedance (10 Ω) was selected to connect the wire to the DUT to represent a moderately poor connection to the enclosure. The radiated emissions from the two- and three-wire setups with differential-mode excitation are shown in Figs. 21 and 22, respectively.

When the ends of the wires on the termination side are open or shorted to the ground plane, the structures are symmetric. In this case, there is little common-mode current flowing in the circuit and very low radiation is observed. Large resonances are shown with open or short terminations because the source impedance is 0 Ω and there is little loss in the system. Resonances would be damped in this case with a larger source impedance. When using imbalanced terminations, there is noticeable differential to common-mode conversion for both the two- or three-wire case, which significantly increases the radiated emissions. As shown in Figs. 21 and 22, the results using cosimulation and using the standard EM simulation with ideal terminations are nearly identical. The difference between the radiation evaluated using the measured S-parameters from the prototype and using ideal terminations was less than a 3.2 dB up to 200 MHz (red curves in Figs. 21 and 22).

C. Tertiary-Mode Excitation With Imbalanced VHF LISN

balanced terminations for the three-wire setup. The excitation is 1 V with zero output impedance. The radiated emissions with a tertiary-mode excitation are shown in Fig. 24. When the wires are open or short, the structure creates a highly resonant system. In the presence of terminations, these resonances will be dampened, regardless of their source. The radiation of the prototype is very close to the radiation with an ideal termination, with less than 3-dB error up to 200 MHz (red curve in Fig. 24).

D. Impact of Termination Condition on Measurement Uncertainty of Imbalanced VHF LISN

The results of radiated emission measurements are affected by the uncertainties listed in [20] and [21]. This section investigates the impact and degree of influence of mains cable termination conditions on the standards compliance uncertainty (SCU) [20], [21]. The SCU is dependent on termination conditions over the frequency range where power cable radiation dominates [20]. Considering the tolerance of the termination impedance, the variation of the emission levels can be calculated, which also allows calculation of the measurement uncertainty influence [20].

The sensitivity of the radiation behavior of the LISN to deviations in the magnitude and phase of the terminating impedance from the ideal case should be analyzed to understand their impact on the SCU. Uncertainty is considered in the CISPR 16-4-1 standard [21]. Here, the average combined standard uncertainty (Uc-scu), including the terminating condition of the main cable, is defined to be

where CISPR/TR 16-4-1 specifies [19], [20] the following.

1) Uc,MIU, combined measurement instrumentation uncertainty, of 2.5 dB.

2) Ua, uncertainty from the main cable arrangement, of 3.5 dB. This value will depend on the termination conditions.

3) Uc, uncertainty in the operating condition of DUT, of 1.7 dB.

4) Ub, the uncertainty in termination conditions.

Assuming a rectangular probability distribution for the uncertainty of the cable terminating conditions, which is considered in CISPR 16-4-1 [21], the uncertainty in the terminating condition is given by [20]

where Emax and Emin are the maximum and minimum electric field strengths in dB µ V/m, respectively. If we consider the maximum deviation due to the termination condition of the prototype up to 200 MHz to be 3.5 dB (see Fig. 22 for a three-wire termination with DM excitation), the uncertainty in the terminating condition Ub is only 1 dB. Using (2), the average combined standard uncertainty (Uc-scu) is 4.7 dB. The expanded standard uncertainty Uscu, VHF-LISN is [20]

Compared to the 50-Ω LISN in [20] with an expanded standard uncertainty Uscu, VHF-LISN = 12˜dB, the expanded standard uncertainty of the imbalanced two- or three-wire VHF LISN has been improved by about 2.5 dB. Compared to the 15.5-dB uncertainty defined in CISPR 16-4-1 [21], the SCU for the imbalanced two- or three-wire VHF LISN is improved by about 6 dB. It should be noted that only a single device was studied here, and results may change with other devices.


The effect of the differential- to common-mode conversion was investigated using a power line communication device. The goal was to investigate the impact of termination conditions on emissions with a DUT that uses a strong differential-mode signal to transmit data over power lines [11], [12]. The test used different termination conditions, a balanced LISN, an imbalanced LISN, and no LISN. A balanced LISN (see Fig. 1) was prototyped to compare with the imbalanced LISN. The balanced LISN had a 50-Ω impedance on each line with less than 1.5-Ω variations in magnitude and less than 5◦ variations in phase over the frequency range from 30 to 200 MHz.

A block diagram of the measurement setup inside the semianechoic chamber is shown in Fig. 25. The two DUTs are HD power line adaptors (PLA5456), which communicate with each other through power lines. The measurement setup is shown in Fig. 26. The DUTs and LISN are mounted on the floor such that the power cable produces a loop with the maximum radiated emissions toward the antenna. A personal computer (PC) was needed to communicate with the DUT. To generate the highest differential-mode current, the DUT was operated under its maximum data rate. The measurement was performed for both horizontal and vertical polarizations and with 1- to 4-m scan heights of the antenna. The table was also partially rotated to capture the maximum radiation.

The goal was to show the differential- to common-mode conversion by using an imbalanced LISN, but also to show the impact of connecting the LISN to different power nets, as well as to show the impact of the power nets on emissions when no LISN was used. Connecting the LISN to different power nets helps to show if it has reproducibility issues. Different power networks were created by adding a soldering iron, a linear dc power supply, different wires with terminations, such as 2 nF, power cords, and strip lines, to the outlet inside the chamber.

Changing the power net before the LISN should have no effect, as the LISN isolates well. With different termination impedances, however, the radiation should change noticeably if no LISN is used. While different emissions are expected from a balanced or imbalanced LISN, both are expected to generate somewhat stable curves because both LISNs isolate the DUT from the power net. Higher radiation is expected using an imbalanced LISN rather than by using a balanced LISN, since the imbalanced LISN converts differential-mode current into common-mode current. Radiation results for all power networks and using three different termination conditions (balanced LISN, imbalanced LISN, and no LISN) are shown in Fig. 27. When both power line communication devices are ON, the broadband signal below 80 MHz is representative of the data transfer from the DUT. The DUT has no differential-mode energy above 80 MHz. Some observations are (a) when not using an LISN, the radiated emissions vary by up to 12 dB, since the termination is not controlled. (b) with a balanced or an imbalanced LISN, radiation has less than 3-dB variation, because the LISNs isolate the network and provide a well-defined termination. (c) the conversion with the imbalanced LISN is as high as 12 dB. (d) the imbalanced LISN generates emissions that are close to the worst of the measured emissions when connecting directly to the power networks, and (e) both LISNs have no reproducibility issues.

Neither impedance, nor conversion, has been characterized for the chamber used for measurement. Therefore, the observed variation in radiated emission is expected to be large ΔE > 10 dB. However, both balanced and imbalanced LISNs have controlled terminations and will not cause reproducibility problems because the data have less than 3-dB variation for different power networks. In general, the data show that for a device that has a strong differential-mode current, the imbalanced LISN brings the emissions to a more realistic level. The result from the balanced LISN ignores the differential- to common-mode conversion and gives unrealistically lower emission.

The conversion of course should correlate to the currents on the wires. The common-mode and differential-mode currents have been measured with a F65 current clamp at a few points along the cables. The maximum current has been captured and the conversion was calculated as the difference between the maximum common-mode current when the cable was terminated with balanced and imbalanced LISNs.

Similarly, the radiated emission conversion was calculated as the difference between the maximum radiated emission when the cable was terminated with balanced and imbalanced LISNs. Fig. 28 shows differential-mode to common-mode conversion, which is calculated from both current (blue curve) and radiated emission (red curve) measurement using balanced and imbalanced LISN. As shown in Fig. 28, the conversion calculated from radiated emission and the common-mode current are quite similar, i.e., this plot validates the conversion ratio with two different quantities, e.g., current, and radiated emissions. Above 80 MHz, the observed differences between the common-mode current for both balanced and imbalanced LISNs are almost zero because the DUT has no transmit energy. At the lower frequency band, the conversion is as high as 12 dB for both radiated emission and common-mode current.


Imbalanced LISNs can be designed for different target impedances. The LISN analyzed in this article was designed for a 150-Ω common-mode impedance, as suggested in the literature [10]–[12]. This impedance can strongly attenuate cable resonances [13]. Since actual power networks show strong resonances, one would underestimate the actual radiation at these frequencies. An alternative would be to design the LISN for a 25-Ω common-mode impedance. While this smaller impedance would strongly increase the radiation at resonances, there is no certainty that the resonances would occur at the same frequencies in the real installation since the actual power line impedances are unknown. A far-reaching design of an LISN might allow one to adjust the common-mode impedance along the Smith chart to identify the worst-case common-mode impedance for a given DUT. The impact of common-mode impedance on the radiation at resonances should be investigated in future.

Alternatively, it may be worthwhile to further study the PE line impedance in a variety of application areas, for example, in an urban area or in light industry area, and then adjust the LISN impedance according to the application.


An analysis of an imbalanced two- or three-wire VHF LISN was conducted in terms of its mode conversion and termination impedance. It was demonstrated that an imbalanced termination impedance provides a specified degree of conversion from differential to common mode, which can lead to more representative radiated emission test results. To ensure spectral emission control, an imbalanced LISN is needed. An imbalanced two- or three-wire VHF LISN was prototyped. The impedances in the prototype had less than 10% error in magnitude and 30◦ in phase compared to the impedances for an ideal imbalanced LISN, as specified by the working group. A 3-D full-wave simulation was performed to investigate the maximum radiation of a twoor three-wire setups using an imbalanced termination. It was demonstrated that the performance of the prototype leads to less than 3.5-dB error as compared to an ideal imbalanced LISN. In EMC applications, this error threshold is acceptable. It is therefore suggested that the impedance of an imbalanced VHF LISN vary by less than ± 30◦ in phase and ± 10%. For the main cable termination, the standard compliance uncertainty has been considered in CISPR 16-4-1 to be 15.5 dB. This uncertainty has been improved to about 9.5 dB for the proposed prototype. The differential- to common-mode conversion for an imbalanced termination was measured with two power line communication device to be as high as 12 dB considering both current and radiated emissions. Using different power nets inside an anechoic chamber, it was demonstrated that the chamber-tochamber reproducibility will be much better if an imbalanced LISN is used in every chamber.

Posted on

PB2021.10 1 Ω Disk Resistor Full-Wave Modeling for JS-002 Standard

Download PDF – 1 Ω Disk Resistor Full-Wave Modeling for JS-002 Standard

Abstract –A 1 Ω disk resistor is specified in the CDM standard as the current sensing element. However, its transfer impedance is frequency dependent which is not considered in the standard. In this work, a full-wave model and a simple equivalent circuit model is provided to explain the root cause of the variation the transfer impedance of the 1 Ω disk resistor.

I. Introduction

When the pin or pad of a charged IC approaches an external metal object, and the breakdown voltage is exceeded a Charged Device Model (CDM) event occurs. During CDM testing, the discharge current of the CDM event is measured by a 1 Ω disk resistor sensing element that is located at the top of a pogo pin probe as described in the industry standard ANSI/ESDA/JEDEC JS-002 [1]-[3]. The resistance of this element is specified to have a value of 1.0 Ω ± 10% and a transfer impedance that does not have a deviation relative to the DC value greater than 3 dB up to 9 GHz [3]. The standard does not consider frequency variations of the transfer impedance when calculating the current from the measured voltage. This work investigates this assumption by characterizing the sensing element (1 Ω disk resistor) of a field induced CDM tester in the frequency domain. Additionally, a simple circuit model and a full-wave model are presented to explain the variation of the transfer impedance up to 27 GHz.

II. CDM tester

A. Discharge Circuit

The cross section of a CDM tester [3] is shown in Figure 1.

To charge the device, the field plate is brought to the specified charge voltage, then the pogo pin is lowered to contact the DUT pin. A spark is initiated and the current flows via the pogo pin to the 1 Ω disk resistor (Figure 2). The transfer impedance, properties of the oscilloscope and cable losses will determine the voltage that is displayed at the scope. The voltage is measured across the 1 Ω disk resistor from which the discharge current waveform is then calculated. However, the transfer impedance of the 1 Ω disk resistor is not constant over the frequency range and deviates from the DC value of the 1 Ω resistor [5].

B. 1 Ω Disk Resistor Measurement

As shown in Figure 2, the disk resistor consists of a resistive sheet on one side and a ceramic substrate made from beryllium oxide (BeO) on the other side that provides mechanical strength. In a CDM test head, the resistive sheet can be mounted downward (Figure 3a, resistive sheet towards pogo tip) or upward (Figure 3b, resistive sheet towards coaxial line).

Figure 4a shows a fixture to measure the disk resistor. The fixture is composed of two 50 Ω surface mountable connectors and a plate that aligns the disk resistor. The thickness of the plate was chosen to prevent any gap between the 50 Ω connectors and the surface of the disk resistor. The surface mount connectors were connected to ports 1 and 2 of a VNA and port extensions were performed up to the connector surfaces (Figure 4a). Figure 4b shows the definition of current and voltage of the two-port measurement setup (disk resistor setup measured with VNA). Knowing the S-parameter across the disk resistor, the transfer impedance of the disk can be calculated as in [6]:

As shown in Figure 5, the S21 and S12 of the disk are nearly identical which gives evidence for the accuracy of the measurement. However, the S22 and S11 are different (Figure 6). Port 1 of the measurement fixture (Figure 4) is toward the resistive sheet of the disk resistor, and port 2 is connected to the ceramic side of the disk. The reflection coefficient S11 is nearly flat but S22 has lower value at higher frequencies. The difference between S11 and S22 is important in understanding the CDM system behavior at high frequencies. Ringing was observed in the time domain discharge waveform if the substrate is mounted toward the pogo-pin but it reduced if the disk was flipped. The ringing will be discussed in Section IV of this paper. At low frequencies, the transfer impedance equals the disk resistance. Thus, the impedance measured with VNA should match the measurements obtained by an LCR meter (1 Ω @ 1kHz).

As shown in Table I, both LCR Meter (@ 1kHz) and disk measurements (Figure 4) show about 1 Ω for different orientations of disk (up to 1GHz). For each sample, both methods showed nearly identical low frequency values (Figure 7). At higher frequencies, the transfer impedance obtained using (1) is increasing with frequency (Figure 8) for all disks. Additionally, manufacturing tolerances lead to variations between disk samples. The behavior of the S21 can be explained using a simple circuit model in the next section.

III. Device Modeling

A. Simple Circuit Model

Although different disk resistors showed slightly different transfer impedance values versus frequency, they all behave similar (Figure 8). The transfer impedance of the disks increased with frequency from 1 Ω @ 1 MHz to about 3 Ω @ 20 GHz. This behavior can be explained by the influence of the ceramic substrate. It forms a short, 13 ps delay transmission line (see Figure 2) which acts as transmission line transformer and changes the match of the 1 Ω disk side to the 50 Ω system. The substrate has a relative permittivity of around 7. However, the structure is not easy to model in 3D due to possibility of higher order modes if they are excited. Only considering TEM modes, a simple circuit model is created in Advanced Design System (ADS) [7] to evaluate the influence of the short ceramic transmission line (Figure 9).

Based on the geometry and permittivity of the Beryllium Oxide (BeO) substrate, the characteristic impedance of the ceramic portion was calculated to be roughly 17 Ω. A 13 ps long 17 Ω lambda/4 transmission line transformer converts 50 Ω to transfer impedance of about 3 Ω at 20 GHz which can be calculated with (1).

This transmission line behavior explains the observed increase of S21 and transfer impedance. As shown in Figure 10, S11 is almost flat over the entire frequency range whereas S22 decreases with frequency as the short transmission line changes the match to 50 Ω. A comparison between measured impedance and calculated from the simple circuit model will be discussed in the next section.

B. Full-wave Model

The simple circuit model gives a qualitative insight into only the dominating effects, excluding the influence of the skin effect, higher order modes, and details of the geometry. A full-wave model can reveal additional details about the frequency-dependent impedance. Figure 11 shows the core elements of the full-wave model: the 50 Ω connectors, the geometry of the disk resistor, and two waveguide ports which are placed across the 50 Ω connectors. Two short 50 Ω connectors are placed on both sides of the disk. As shown in Figure 2, the 1 Ω disk resistor has a resistive sheet on one side and a ceramic carrier made of beryllium oxide on the other side. A thin resistive sheet material (this does not model skin effect) and BeO were imported from the library of CST Studio Suite [8].

C. Comparison Between Measurement and Simulation

Figure 12 and Figure 13 compares the magnitude and phase variation data for the measured, circuit model, and full-wave model of the 1 Ω disk resistor. Both models and the data agree on the increase of the impedance above 1 GHz and the peak around 20 GHz. This increase and the peaking may cause some error in the peak current measurement for CDM if the actual current contains relevant spectral content in this frequency range.

The fact that the simple circuit model and the measured data match up to 20 GHz can be seen as indicator that the short transmission line is the dominating reason for the observed increase in transfer impedance and the behavior of S11 and S22. The full-wave model predicted a higher peak value, 3.3 Ω relative to 2.6 Ω in the measurements (Figure 12 for resistive sheet down). The reason is not known, but as the frequency matches the other data one can be assured that the dielectric constant of the BeO was correctly estimated from literature data. The full-wave simulation shows additional resonant behavior around 25 GHz which is also seen in the measurements in the same frequency range. We have not investigated the field distribution at these frequencies within the full-wave results to identify the nature of these resonances. Furthermore, the transfer impedance of the disk may decrease at higher frequencies when skin effect starts to decouple the front and the back side of the very thin resistive sheet. As the authors do not know the exact thickness and material of the resistive layer, it is not known above which frequency the decoupling effect of the skin effect would reduce S21. The data indicates that this is not the case below 20 GHz, as the simple ADS model matches the measurement in its principle behavior. Our full-wave model is also not able to simulate skin effect as an infinitely thin electrical layer was used to model the resistive sheet. The metallization of the resistive layer may not be fully homogeneous. This would cause a current flow that is not radially symmetric. As known from current shunts, this would increase the mutual inductance between both sides of the resistive disk. Such a behavior is not observed which leads to a tentative conclusion that the resistive sheet is homogeneous within the boundaries of the analysis.

IV. Effect of Disk Orientation on CDM Event

To investigate the effect of disk orientation, both measured and simulated results have been compared in a CDM test setup.

A. Effect of Disk Orientation on Measured Discharge Current

CDM classification levels have been reduced [9] and that further reductions are to be expected. This will lead to a faster rise time in CDM. This, paired with faster I/O on ICs may lead to measurement problems in CDM testing due to the mounting direction of the disk resistor. To investigate the effect of disk orientation, CDM discharge tests have been measured using a 23 GHz bandwidth oscilloscope [10] with different orientations of disk resistor as shown in Figure 3. One disk resistor was used within one test head by flipping the orientation between tests to prevent any unwanted effects or variation in the test setup. Discharge data from multiple pogo-pins with different length and discharge currents have been captured for charge voltage of 500 V (Figure 14 through Figure 17).

As shown, all plots have a low frequency component around 1 GHz which is the main CDM discharge current. However, there are some high frequency components as well which create ringing waveform over the low frequency waveform. As shown, high frequency ringing was observed in discharge current when the resistive sheet of disk resistor was mounted upward (Figure 3b).

However, the ringing is weaker if the resistive sheet of disk resistor is mounted downward (substrate toward oscilloscope and the resistive sheet toward DUT as shown in Figure 3a). To isolate the ringing from the familiar low frequency response, a Maximum Overlap Discrete Wavelet Transform based Multiresolution Analysis (MODWT MRA) was used [11]. This method yields excellent decomposition and reconstruction while maintaining sharp edge definition and minimizing non-causality introduced by traditional high pass filtering. The high frequency ringing signal was found to be well isolated from the rest of the signal by using the db7 wavelet with a scaling factor of 2. Furthermore, the Wigner-Ville distribution [12] of ringing is used to visualize the time dependent frequency composition of the time dependent current.

The time domain signal, the power spectral density and time-frequency scalogram of ringing for different pogo-pins and for resistive sheet up and down is shown in Figure 18 and Figure 19. As indicated in timefrequency spectra of Figure 18, when the disk resistor is mounted upward for pogo pin length of 8.25 mm, 9.4 mm and 10.5 mm, there are two main high frequency component which make up the ringing with the corresponding interference term between the two main components.

However, in all time-frequency spectra of Figure 19, there is only one frequency component. Similarly, when the resistive sheet of disk resistor is mounted upward the power spectrum has two main frequency components for pogo pin length of 8.25 mm, 9.4 mm and 10.5 mm. Table II summarizes the two observed frequencies if the resistive sheet of disk is mounted upward (Figure 18).

Two sinusoidal signals are used in Figure 20 to reconstruct the ringing for pogo pin of 10.5 mm (blue curve in Figure 20) and is compared with the original ringing (red curve in Figure 20).

Therefore, it is possible to reconstruct the ringing by summing two sinusoidal signals e.g., f1 and f2 as shown in Figure 20. This motivates us to consider the nature of ringing and determine the physical agents which correspond to these responses.

Figure 21 shows the half wavelength versus frequency (blue curve) and is compared with the length of the pogo pins versus the first sinusoidal signal (f1) from Table II (black curve). As shown, the first sinusoidal signal (f1) is directly related to the length of pogo pin and can be calculated relative to the length of the pogo pin. The second sinusoidal component is related to the disk orientation. As shown in Figure 18 and 19, the second sinusoidal signal (f2) exist for resistive sheet upward for pogo pin 8.25 mm, 9.4 mm and 10.5 mm. However, this signal disappears when the resistive sheet is mounted downward. For the pogo pin 6.6 mm, two sinusoidal signals cannot be distinguished since f1 is very close to f2. When the resistive sheet of disk resistor in mounted downward (resistive sheet toward DUT), only one frequency can be observed in time-frequency spectrum of ringing signal (Figure 19) which indicate the effect of disk orientation.

B. Effect of Disk Orientation in Simulation

As shown in Figure 18 through Figure 20, two sinusoidal signals contribute to the high-frequency ringing of the CDM discharge current. A full-wave model is created for CDM test setup (Figure 22) to obtain a qualitative insight into the dominating effects up to 30 GHz. Two discrete ports are placed on both sides of the pogo pin providing the corresponding connection for co-simulation simulation in ADS which is shown in Figure 23. Measured S-parameters of the 1 Ω disk resistor from Section II or simulated S-parameter file from Section III can be imported into ADS model of Figure 23. It is also possible to use the simple circuit model of the disk resistor (Figure 9) into the circuit model of Figure 23. As shown in Figure 24 and Figure 25, if the orientation of the disk is changed, the high frequency ringing also changes. The first ringing in the discharge current relates to the length of the pogo pin which exists in the discharge current regardless of disk orientation. However, the second ringing corresponds to the disk orientation and will disappear if the resistive sheet of the disk is mounted toward the DUT (red curve of Figure 24 and Figure 25). If the substrate of the disk is mounted toward the DUT (blue curve in Figure 24 and Figure 25), the waveform has more ringing (high frequency contents).

The effect of disk orientation in simulated data (Figure 24 and Figure 25) is not as strong as the measured waveform (Figure 14 through Figure 17), but in principle they follow the same behavior, i.e., ringing is stronger if the ceramic substrate is mounted toward the DUT and gets weak if the resistive sheet is mounted toward the DUT.

It is known that the current measured at disk is not necessarily equal to the current at the DUT [13]. Ultimately, the current at the DUT is the stress that the device experienced. For accurate comparison between measured and simulated data in this paper, only current at the disk resistor was studied. Future work should incorporate current at the DUT to get a more accurate measurement of the stress that the device experiences.

A. Discussion

The fundamental question is how important is the frequency response of the transfer impedance of the disk resistor above 10 GHz?

Present ICs have data rates up to 50 GHz and more. Thus, their I/O can be damaged by high frequency content of every strong signal [14]. The charge voltages of CDM will be further reduced such that the rise times will further reduce [9]. Thus, the importance of the larger than 10 GHz spectral content will increase. Right now, CC-TLP testers can be based on a 40 ps or less transmission line pulser [14]. A 30 ps rise time equates to 10 GHz. To avoid problems in testing of future ICs and for comparing test methods, the analysis > 10 GHz is suggested.

V. Conclusion and Further Investigations

The frequency response of the transfer impedance of a 1 Ω disk resistor has been investigated through measurement, full-wave modeling, and a simplified equivalent circuit. The transfer impedance increases with frequency and shows a maximum of about 3 Ω at 20 GHz for disks mounted downward (resistive sheet toward DUT). However, the transfer impedance increases with frequency and shows a maximum of about 15-20 Ω around 20 GHz for a disk mounted upward (resistive sheet toward oscilloscope).

This is explained by considering the inner structure of the 1 Ω disk resistor. Only one side of the resistor’s ceramic carrier contains the resistive sheet material. Thus, it is asymmetric. The thin ceramic carrier creates a short transmission line. The effect of this short transmission line section is clearly visible in measurement, simplified and full-wave simulation both in S11, S22 and S21 data. It cannot match the 1 Ω but it strongly changes the match and causes an increase of the transfer impedance. The CDM current has been captured with different pogo pin lengths. High frequency ringing was observed. It can be explained as the sum of two sinusoids. The first sinusoidal signal was directly related to the resonance frequency of the pogo pin structure, i.e., its length and the second one was created due to the transfer impedance of the disk orientation.

VI. Acknowledgements

We would like to thank Dr. David Johnnsson and Dr. Timothy Maloney for their useful discussions and comments.